This invention relates to a dc-to-dc converter capable of adaptation to variations in load or input voltage.
As disclosed for example in the U.S. Pat. No. 5,719,755, a typical prior art comprises a direct-current power supply, a transformer having a primary, a secondary and a tertiary winding, a switching device, a rectifying and smoothing circuit, and a control circuit. Connected across the dc power supply via the transformer primary, the switching device is turned on and off under the direction of the control circuit.
The rectifying and smoothing circuit may be either of two different types. The first type comprises a rectifying diode and a capacitor. The rectifying diode is so connected to the transformer as to be reverse-biased by a voltage induced in the transformer secondary during the conducting periods of the switching device connected to the transformer primary, and forward-biased by a voltage inducted in the transformer secondary during the nonconducting periods of the switching device. The output voltage of the rectifying diode is smoothed by the capacitor.
The second type of rectifying and smoothing circuit has a rectifying diode that is so connected to the transformer secondary as to be forward-biased by the voltage induced in the transformer secondary during the conducting periods of the switching device. The output of this rectifying diode is connected to a choke coil, thence to a smoothing capacitor, and a diode is provided to form a closed circuit in combination with the choke coil and the smoothing capacitor.
The voltage regulator with the first type of rectifying and smoothing capacitor is commonly referred to as the flyback or reverse switching regulator, and that with the second type as the forward switching regulator.
There are strong demands for dc-to-dc converters of higher operating efficiency. Improving the efficiency of dc-to-dc converter depends to a large measure upon reduction of the power loss of the switching device. To this end the U.S. Patent cited above employs what is termed a quasiresonant capacitor, which is connected in parallel with the switching device. Connected in parallel with the switching device, the capacitor is gradually charged while the switching device is off, causing a gradual rise in the voltage across the capacitor and across the switching device. With use of a bipolar transistor or a field-effect transistor as the switching device, current will continue flowing therethrough after it is turned off, due to the carrier storage of the semiconductor. With the provision of the resonant capacitor as above, however, the voltage across the switching de vice does not rise too sharply after it is turned off. The result is the reduction of the switching loss, or of the loss of power equivalent to the product of the current through, and the voltage across, the switching device. Also reduced is the voltage surge or the noise when the switching device is turned off.
Preparatory to causing conduction through the switching devices the voltage across the same is gradually reduced by the resonance of the inductance of the transformer primary and the capacitance of the capacitor connected in parallel with the switching device. The switching device is turned on when that voltage is reduced to zero or thereabouts. Thus is accomplished the zero-voltage switching of the switching device, with consequent reduction of the switching loss.
In such a quasiresonant switching regulator, incorporating means for holding the output voltage constant, variations in the voltage requirement of the load manifest themselves as changes in the on-off rate (herein-after referred to as the switching frequency) of the switching device. A drop in the voltage requirement of the load, for instance, results in an increase in switching frequency. A higher switching frequency means that the switching device is actuated a greater number of times per unit length of time. Since the switching device causes a loss each time it is actuated, a loss per unit length of time also increases as the switching device is actuated oftener per unit length of time. Consequently, despite use of the quasiresonant capacitor, the efficiency of the noted prior art switching regulator did not necessarily improve.
It has been known to set a limit upon the switching frequency during operation under a light load, as taught for instance by Japanese Unexamined Patent Publication No. 8-289543. This objective has so far been attained by compulsorily imposing a lower limit on the nonconducting periods of the switching device. The actual nonconducting periods of the switching device are not permitted to fall short of the mandatory minimal nonconducting period thus imposed.
In a dc-to-dc converter with the predetermined minimum required nonconducting period, in the event of a substantive drop in the voltage requirement of the load, the switching device is not immediately turned on, but upon lapse of the predetermined minimum nonconducting period, upon completion of the production of the flyback voltage due to the release of the energy that has been stored on the transformer during the preceding conducting period of the switching device. With the completion of the production of the flyback voltage during the minimum nonconducting period, so-called ringing will occur due to the inductance of the transformer winding and the parasitic capacitance or resonant capacitance of the switching device. The switching device is turned on in the course of this ringing. The voltage across the switching device may be high due to the ringing at the end of the minimum nonconducting period, so that the switching device is turned on when the voltage across the same grows sufficiently low after the expiration of the minimum nonconducting period. This known method of controlling the switching device succeeded in material curtailment of the switcing loss.
It has later proved, however, with the fixed minimum nonconducting period, as has been the case heretofore, the switching frequency has tended to become unstable in the event of fluctuations in the input voltage or in the voltage requirement of the load. Let us consider the case in which the voltage requirement of the load changes from a first, relatively heavy state, such that the flyback voltage is generated longer than the minimum nonconducting period, to a second, relatively light state in which the flyback voltage is generated shorter than the minimum nonconducting period. The instability of the switching frequency has occurred just when the duration of the flyback voltage becomes less than the minimum non-conducting period.
The foregoing discussion of instability in switching frequency will be better understood from a consideration of FIGS. 6 and 7. The indicia V1 in these figures denotes the voltage across the switching device, the voltage being due to the transformer flyback voltage and ringing voltage. The indicia T1, at V13 in FIG. 6 denotes the predetermined minimum non-conducting period.
Under a relatively heavy load, as represented by FIG. 6, the duration T0 of the flyback voltage is longer than the minimum nonconducting period T1. The switching device will then be turned on immediately upon expiration of the flyback voltage, resulting in the continuation of the known self-excited oscillation. Then, with a gradual lessening of the load, the conducting periods Ton of the switching device will grow less, and so will the durations T0 of the flyback voltage, until at last the flyback voltage duration becomes shorter than the minimum nonconducting period T1.
As will be understood from FIG. 7, the switcing device will be inhibited from turning on at the end of the duration T0 of the flyback voltage when the flyback voltage duration becomes less than the minimum nonconducting period T1, as above. The switching device will be turned on when the voltage across the same becomes approximately zero after the end of the minimum nonconducting period T1. The nonconducting period of the switching device will increase if the device is held turned off until the voltage across the same becomes approximately zero, instead of until the end of the minimum nonconducting period. The result will be a drop in the output voltage.
The dc-to-dc converter under consideration is usually equipped to keep its output voltage constant. Therefore, in order to compensate for the voltage drop, the conducting period of the switching device will become longer. The duration T0 of the flyback voltage will grow longer in proportion with the conducting period of the switching device, until the flyback voltage duration exceeds the minimum conducting period again. The nonconducting period of the switching device will then be not limited to the minimum nonconducting period T1. With the longer conducting period, however, the resulting rise in output voltage will cause a decrease again in the conducting period, until the flyback voltage duration T0 again becomes less than the minimum nonconducting period T1. Then the non-conducting period will again be under the limitation of the minimum non-conducting period T1.
With the nonconducting periods of the switching device limited as above according to the prior art, the conducting and nonconducting periods of the switching device changed cyclically, resulting in instability in switching frequency. The switching frequency instability resulted in turn in variation in the frequency of the noise generated by the switching device. The variable frequency noise was difficult of suppression. The switching frequency instability also caused noise production by the transformer due to magnetostriction, and instability in the on-off control of the switching device.
In view of the foregoing state of the above, the present invention has it as an object to provide a dc-to-dc converter that is improved in the efficiency of operation under a light load, in stability of operation, and in noise production.
The present invention will be briefly explained using the reference characters seen in the attached drawings showing the embodiments of the invention. The reference characters, however, are meant purely for a better understanding of the invention and should not be taken in a limitative sense.
The dc-to-dc converter according to the invention, for delivering direct-current power to a load 26, comprises a dc power supply 1 for providing a unidirectional voltage, a switching device 3 connected between the terminals 18 and 19 of the power supply 1 and having a first and a second main terminal and a control terminal, inductance means 2 or 2a connected in series with the switching device and adapted to store energy during the conducting periods of the switching device and to release the energy during the nonconducting periods of the switching device, a rectifying and smoothing circuit 6 connected to the inductance means, output voltage detector means 8 and 10 for detecting a signal indicative of the output voltage of the rectifying and smoothing circuit, switch voltage detector means 11 or 11a or 11b for providing a signal indicative of a voltage between the first and the second main terminal of the switching device, and switch control means 13.
The switch control means 13 is connected to the output voltage detector means 8 and 10 and the switch voltage detector means 11 or 11a or 11b and the switching device 3 for producing a switch control signal for on-off control of the switching device 3 and for applying the switch control signal to the control terminal of the switching device. The switch control means 13 performs the functions of:
1. Determining the conducting periods Ton, of the switching device 3 so as to keep constant the output voltage in response to the output of the output voltage detector means 8 and 10.
2. Forming a signal indicative of a first minimum nonconducting period T1. to which the nonconducting periods Toff of the switching device are to be limited.
3. Forming a signal indicative of a second minimum nonconducting period T2 longer than the first minimum nonconducting period T1.
4. Selectively providing the first and the second minimum nonconducting period signals.
5. Detecting the duration T0 of the flyback voltage from the inductance means 2 or 2a. 
6. Judging whether the flyback voltage duration T0 is shorter than the first minimum nonconducting period T1 or not.
7. Judging whether the flyback voltage duration T0 is longer than the second minimum nonconducting period T2 or not.
8. Limiting the nonconducting periods Toff of the switching device 3 under the second minimum nonconducting period T2 when the flyback voltage duration T0 proves to be shorter than the first minimum nonconducting period T1.
9. Limiting the nonconducting periods Toff of the switching device 3 under the first minimum nonconducting period T1 when the flyback voltage duration T0 proves to be longer than the second minimum nonconducting period T2.
10. Terminating the nonconducting periods Toff of the switching device 3 when the signal indicative of the voltage of the switching device, obtained by the switch voltage detector means 11 or 11a or 11b upon termination of the first T1 or second T2 minimum nonconducting period, grows equal to or less than a predetermined reference value Vr1 or Vr2xe2x80x2.
As indicated in claim 2, the switch control means 13 preferably comprises switch control signal forming means 46, 46a, 47, 50 or 50a, a minimum nonconducting period signal generator circuit 73, 120 or 120xe2x80x2, flyback voltage duration detector means 101 or 130, and judgment means 102 or 133. The switch control signal forming means 46, 46a, 47, 50 or 50a is connected to the output voltage detector means 8 or 10 and the switch voltage detector means 11, 11a or 11b in order to form the switch control signal for on-off control of the switching device 3. The functions of the switch control signal forming means 46, 46a, 47, 50 or 50a include:
1. Determining the conducting periods Ton of the switching device 3 so as to keep the output voltage constant in response to the output from the output voltage detector means 8 and 10.
2. Terminating the nonconducting periods Toff of the switching device 3 when the signal indicative of the voltage across the switching device, obtained from the switch voltage detector means 11, 11a or 11b, grows equal to or less than the predetermined reference value Vr1 or VR2xe2x80x2.
The minimum nonconducting period signal generator circuit 73, 120 or 120xe2x80x2 selectively puts out a signal indicative of the first minimum nonconducting period Ti, and a signal indicative of the second minimum nonconducting period T2, which is longer than the first period T1, for limiting the nonconducting periods Toff of the switching device 3. These first and second minimum nonconducting period signals are delivered to the switch control signal forming means.
The flyback voltage duration detector means 101 or 130 detects the duration To of the flyback voltage due to the inductance means 2 or 2a. 
The judgment means 102 or 133 is connected to the minimum nonconducting period signal generator circuit and the flyback voltage detector means. The functions of the judgment means 102 or 133 include:
1. Ascertaining whether the flyback voltage duration T0, detected by the flyback voltage duration detector means 101 or 130 is shorter than the first minimum nonconducting period T1 or not.
2. Ascertaining whether the flyback voltage duration T0 is longer than the second minimum nonconducting period T2 or not.
3. Causing the minimum nonconducting period signal generator circuit 73, 120 or 120xe2x80x2 to deliver the signal indicative of the second minimum nonconducting period T2 to the switch control signal forming means when the flyback voltage duration T0 proves to be shorter than the first minimum nonconducting period T1.
4. Causing the minimum nonconducting period signal generator circuit 73 or 120 to deliver the signal indicative of the first minimum nonconducting period T1 to the switch control signal forming means when the flyback voltage duration T0 proves to be longer than the second minimum nonconducting period T2.
As set forth in claim 3, the time difference Ta between the first minimum nonconducting period T1 and the second minimum nonconducting period T2 is preferably from 0.1 to 10 microseconds.
As set forth in claim 4, a resonant capacitor should preferably be provided.
As set forth in claim 5, means 12 should preferably be provided for combining the output from the current detector means 4, the output from the output voltage detector means 8, and the output from the switch voltage detector means 11 or 11a. The resulting output from the combining means may be fed into the first and the second comparison means 46 and 47. The switch control means will then be more effectively integrated and reduced in cost.
As set forth in claim 6, initialization signal generator means 51 should preferably be provided. Also, the minimum nonconducting period signal generator circuit 77 should preferably comprise a sawtooth voltage generator circuit 72, a source 91 of a reference voltage for determination of the minimum nonconducting periods, a comparator 92 for minimum nonconducting period determination, and a nonconducting period pulse forming circuit 95.
As set forth in claim 7. the control pulse forming circuit 50, 50a or 50b should preferably comprise a first circuit 71, 71a or 71b and a second circuit 96.
As set forth in claim 8, the first circuit 71, 71a or 71b should preferably comprise a wave-shaping circuit 77 and a flip-flip 78.
As set forth in claims 9 and 10, the first circuit 71 or 71a for termination of the nonconducting periods should preferably comprise two flip-flops.
As set forth in claim 11, the second circuit 98 should preferably take the form of a NOR gate 96a. 
As set forth in claim 12, the minimum nonconducting period pulse forming circuit 95 should preferably comprise an AND gate 97 and a flip-flop 100.
As set forth in claim 13, the flyback voltage duration detector means 101 should preferably be connected to the first circuit 71 and the second circuit 96.
As set forth in claim 14, the reference voltage source 91 for determination of the minimum nonconducting period should preferably take the form of a voltage-dividing circuit.
As set forth in claim 15, the output from the switch voltage detector means 11 may not be input to the combining means 12a for output voltage control.
As set forth in claim 16, a sawtooth voltage V4a may be formed from the output voltage detection signal, and this sawtooth voltage utilized for determination of the moment for terminating the nonconducting periods.
As set forth in claim 17, the inductance means should preferably take the form of a transformer having a primary, a secondary and a tertiary winding 21, 22 and 23.
As set forth in claim 18, the switch voltage detector means 11 may be connected in parallel with the switching device 3.
As set forth in claim 19, the output voltage detector means may be connected to the tertiary winding 23.
As set forth in claim 20, the switch voltage detector means 11 may include a delay capacitor 34.
As set forth in claim 21, the switch voltage detector means 11 may comprise a diode 31 and a resistor 33.
As set forth in claim 22, the inductance means may take the form of a reactor 2a, and the output smoothing capacitor 7 connected in parallel with its winding 21 via a diode 6.
As set forth in claim 23, a maximum nonconducting period may be predetermined.
As set forth in claim 24, the output of the switch voltage detector means 11, 11a or 11b may be connected to a switch, by which control by the output from the switch voltage detector means may be suspended.
As set forth in claim 25, two minimum nonconducting period signal generators 121 and 122 may be provided.
As set forth in claim 26, the flyback voltage duration detector circuit 130 may be connected directly to the inductance means.
As set forth in claim 27, the judgment means may take the form of a phase comparator 133.
The invention as claimed gains the following advantages:
1. With the minimum nonconducting periods T1 or T2 set up, any great decrease in the nonconducting periods Toff of the switching device can be prevented when the load is light. An increase in switching loss per unit length of time is thus restricted, contributing to the higher efficiency of operation under light load.
2. The second minimum nonconducting period T2 will be set up when the duration T0 of the flyback voltage falls short of the first minimum nonconducting period T1 by reason of a drop of the load. Stable switching operation will continue in the face of fluctuations in the load or input voltage as the nonconducting periods of the switching device 3 are fixed at the second minimum nonconducting period T2.
3. The first minimum nonconducting period T1 will be set up when the duration T0 of the flyback voltage exceeds the second minimum nonconducting period T2 because of a rise of the load. In this case, too, stable switching operation will continue as the nonconducting periods of the switching device are fixed at the first minimum nonconducting period T1.
4. In short a smooth transition is accomplished from a first switching state in which the switching device is turned on and off without the limitation of the first minimum nonconducting period because of a relatively great load, to a second switching state in which the switching device is turned on and off under the limitation of the second minimum nonconducting period T2 because of a relatively small load, or vice versa. The switching between the two minimum nonconducting periods T1 and T2 according to the instant invention may be likened to the hysteresis operation of a comparator or Schmidt trigger circuit with a known hysteresis characteristic. With such a smooth transition between the two switching states, not only is the switching frequency stabilized, but also the production of unpredictable frequency noise is precluded, and so is the production of audible noise due to magnetostriction from the inductance means.
The setting of the time difference between the first T1 and the second T2 minimum nonconducting period as in claim 3 will surely provide the desired hysteresis effect.
The provision of the resonant capacitance 5 as in claim 4 will stably lead to resonance, and the zero-voltage turning-off of the switching device 3 will be favorably accomplished, realizing a decrease in switching loss.
The formation of the composite signal as in claim 5 will enable transmission of a plurality of pieces of information through common conductors and terminals, making it easier to integrate the switch control signal forming means.
According to the inventions of claims 6-12, the switch control signal will be formed by simple circuit means.
According to the inventions of claims 13 and 14, the hysteresis operation can be readily effected.
According to the inventions of claims 15 and 16, the control of the output voltage and the detection of the end of each nonconducting period can be implemented independently, so that the required circuitry will be easier of designing.
According to the invention of claim 17, the switching device side and the load side will be more easily separated electrically.
According to the invention of claim 18, the voltage across the switching device will be accurately detected.
According to the invention of claim 19, the output voltage will be easily detected.
According to the inventions of claims 20 and 21, the switch voltage will be easily detected.
According to the invention of claim 22, a high output voltage will be easily obtained by virtue of the reactor.
According to the invention of claim 23, the dc-to-dc converter will be stably started up by virtue of the maximum nonconducting period.
According to the invention of claim 24, the dc-to-dc converter will operate stably when the voltage requirement of the load is very low, as in standby mode, as then the switch control signal is formed which has the maximum nonconducting period.
According to the inventions of claims 2 and 26, the required circuitry will be easier of designing.